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lt4351 1 sn4351 4351fs mosfet diode-or controller n low loss replacement for oring diode in multiple sourced power supplies n external n-channel mosfets for high current capability n internal boost regulator supply for mosfet gate drive n wide input range: 1.2v to 18v n fast switching mosfet gate control n input under and overvoltage detection n status and fault outputs for monitoring n internal mosfet gate clamp n available in a 10-pin msop , ltc and lt are registered trademarks of linear technology corporation. n paralleled power supplies n uninterrupted supplies n high availability systems n n + 1 redundant power supplies the lt4351 creates a near-ideal diode using external single or back-to-back n-channel mosfets. this ideal diode function permits low loss oring of multiple power sources. power sources can easily be ored together to increase total system power and reliability with minimal effect on supply voltage or efficiency. disparate power supplies can be efficiently ored together. the ic monitors the input supply with respect to the load and turns on the mosfet(s) when the input supply is higher. if the mosfets r ds(on) is sufficiently small, the lt4351 will regulate the voltage across the mosfet(s) to 15mv. a status pin indicates the mosfet on state. an internal boost regulator generates the mosfet gate drive voltage. low operating voltage allows for oring of supplies as low as 1.2v. the lt4351 will disable power passage during undervolt- age or overvoltage conditions. these voltages are set by resistive dividers on the uv and ov pins. the undervoltage threshold has user programmable hysteresis. overvolt- age detection is filtered to reduce false triggering. the lt4351 is available in a 10-pin msop. dual 5v redundant supply r load c load gate out v in v dd sw lt4351 uv ov status 1 f 24.9k 1% 232 1% 1.47k 1% 10 f 10 f 5v power supply 1 4.7 h mbr0530 fault gnd si4862dy si4862dy gate out 5v common v in v dd sw lt4351 uv ov status 1 f 24.9k 1% 232 1% 1.47k 1% 4351 ta01 4.7 h mbr0530 fault gnd power supply 2 5v descriptio u features applicatio s u typical applicatio u
lt4351 2 sn4351 4351fs v in voltage ............................................... C 0.3v to 19v out voltage ............................................ C 0.3v to 19v v dd voltage ............................................. C 0.3v to 30v fault, status voltages ........................ C 0.3v to 30v fault, status current ........................................ 8ma uv, ov voltages ........................................ C 0.3v to 9v sw voltage .............................................. C 0.3v to 32v operating temperature range lt4351c .................................................. 0 c to 70 c lt4351i .............................................. C 40 c to 85 c junction temperature (note 2) ............................ 125 c storage temperature range ................ C 65 c to 150 c lead temperature (soldering, 10 sec).................. 300 c order part number ms part marking t jmax = 125 c, q ja = 120 c/w ltzz lta1 lt4351cms LT4351IMS (note 1) absolute axi u rati gs w ww u package/order i for atio uu w consult ltc marketing for parts specified with wider operating temperature ranges. the l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25 c. v in = v out = 5v, v dd = 16.1v, v uv = 0.4v, v ov = 0.2v, gate open, unless otherwise specified. electrical characteristics 1 2 3 4 5 gate v dd v in sw gnd 10 9 8 7 6 out status fault uv ov top view ms package 10-lead plastic msop symbol parameter conditions min typ max units supply and protection v in operating range l 1.2 18 v i vin v in supply current v in = 1.2v, v out = 1.1v, v dd = 12.3v l 1.41 2.0 ma v in = 18v, v out = 17.9v, v dd = 29.1v l 1.71 2.1 ma v uv(th) undervoltage turn-off voltage threshold uv falling l 290 300 310 mv i uv(hyst) i uv hysteresis difference between i uv at v uv(th) + 10mv l 71013 m a and v uv(th) C 10mv i uv uv input bias current v uv = v uv(th) + 10mv l C100 C400 na v ov(th) overvoltage threshold ov rising l 290 300 310 mv i ov ov input bias current v ov = v ov(th) C 10mv l C100 C400 na v f(on) fault pin on voltage i f = 5ma in fault condition l 0.14 0.25 v i f(off) fault pin leakage current v f = 30v, v in = 4.9v l 0.04 1 m a boost supply v br boost regulation trip voltage measured as v dd to v in , rising edge l 10.2 10.7 11.1 v t off boost supply off time 600 ns i swlim boost supply switch current limit l 350 450 650 ma gate drive v ior input to output regulated voltage l 41525 mv d v gl gate voltage limit v in = 5v, v out = 4.9v, v dd = 13v measured with l C2.3 C3 v respect to v dd d v g(max) maximum gate voltage v in = 5v, v out = 4.9v, v dd = 16.1v measured with l 7 7.4 7.8 v respect to v out v goff gate off voltage v out = 5.1v l 0.16 0.30 v i gso gate source current v out = 4.9v, v gate = 9v 0.670 a i gsk gate sink current v out = 4.9v, v gate = 9v 0.670 a
lt4351 3 sn4351 4351fs the l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25 c. v in = v out = 5v, v dd = 16.1v, v uv = 0.4v, v ov = 0.2v, gate open, unless otherwise specified. electrical characteristics note 1: absolute maximum ratings are those values beyond which the life of a device may be impaired. typical perfor a ce characteristics uw undervoltage threshold vs temperature overvoltage threshold vs temperature undervoltage threshold vs v in note 2: t j is calculated from the ambient temperature t a and power dissipation p d according to the following formula: t j = t a + (p d ? 120 c/w) temperature ( c) ?0 290 v uv(th) (mv) 292 296 298 300 310 304 0 50 75 4351 g01 294 306 308 302 ?5 25 100 125 v in = 1.2v v in = 5v v in = 12v v in = 20v temperature ( c) ?0 293 v ov(th) (mv) 297 299 305 0 50 75 4351 g02 295 301 303 ?5 25 100 125 v in = 1.2v v in = 5v v in = 12v v in = 20v v in (v) 0 v uv(th) (mv) 302 306 310 16 4351 g03 298 294 300 304 308 296 292 290 4 2 8 6 12 14 18 10 20 symbol parameter conditions min typ max units v dd operating range l 30 v i vdd v dd supply current v in = 1.2v, v out = 1.1v, v dd = 12.3v, gate open l 3.0 4.0 ma v in = 18v, v out = 17.9v, v dd = 29.1v, gate open l 3.6 5.6 ma status functions d v gis min gate voltage for turning on status v out = 4.9v, i status = 1ma l 0.75 1 v v iogf v in to v out fault voltage v out falling, measured with respect to v in 185 210 230 mv with open gate v ston status pin on voltage i st = 5ma, v out = 4.9v, status on l 0.13 0.25 v i stoff status pin leakage current v st = 30v, status off, v in = 4.9v l 0.04 1 m a specifications are at t a = 25 c unless otherwise specified.
lt4351 4 sn4351 4351fs i vin vs temperature gate off voltage vs temperature gate pin turn on and off waveform with 10nf capacitor load typical sw pin waveform overvoltage threshold vs v in overvoltage hysteresis vs temperature v in (v) 0 v uv(th) (mv) 302 306 310 16 4351 g04 298 294 300 304 308 296 292 290 4 2 8 6 12 14 18 10 20 temperature ( c) 50 ?5 0 ov hysteresis (mv) 10 25 0 50 75 4351 g05 5 20 15 25 100 125 v in = 5v overvoltage turn-off delay vs overvoltage overdrive ov voltage above threshold (mv) 0 0 turn-off delay ( s) 2 6 8 10 20 14 10 20 25 3451 g06 4 16 18 12 5 15 30 35 v in = 5v temperature ( c) ?0 1.0 i vin (ma) 1.1 1.3 1.4 1.5 2.0 1.7 0 50 75 4351 g07 1.2 1.8 1.9 1.6 ?5 25 100 125 v in = 1.2v v in = 5v v in = 12v v in = 20v i vdd vs temperature temperature ( c) ?0 2.0 i vdd (ma) 3.0 3.5 4.5 0 50 75 4351 g08 2.5 4.0 ?5 25 100 125 v in = 1.2v v in = 5v v in = 12v v in = 20v temperature ( c) ?0 0 v goff (v) 0.05 0.15 0.20 0.25 0.50 0.35 0 50 75 4351 g09 0.10 0.40 0.45 0.30 ?5 25 100 125 v in = 5v v out = 5v v in = 5v v out = 5.1v v sw 2v/div v in = 5v 50ns/div 3451 g10 v out = 4.9v to 5.1v square wave turn on turn off v sw 5v/div v in = 5v 500ns/div 3451 g11 l = 4.7 m h typical perfor a ce characteristics uw specifications are at t a = 25 c unless otherwise specified.
lt4351 5 sn4351 4351fs uu u pi fu ctio s gate (pin 1): mosfet gate drive pin. this pin is tied to the gate(s) of the external n-channel mosfet(s). the gate pin drives high when uv is above the v uv(th) threshold, ov is below the v ov(th) threshold and v in is greater than out by 15mv. when not driven high, gate actively pulls to gnd. gate can sink or source up to 600ma. v dd (pin 2): gate drive supply pin. this is the supply pin for the gate drive amplifier. it is either generated by the onboard boost regulator or supplied externally. when turning on the mosfet(s), a large high current pulse flows through this pin. bypass the pin with a 1 m f capacitor placed in close proximity to the part. the voltage on this pin is also the feedback for the boost regulator. if the v dd voltage exceeds the v in voltage by 10.7v, the boost switch is held off. v in (pin 3): input supply pin. this pin is the supply pin for the control circuitry and the boost regulator. it is also one input in conjunction with out for controlling the mosfet(s). bypassing should include a low esr/esl capacitor placed in close proximity to the part. sw (pin 4): boost regulator switch pin. this pin is the boost regulator switch output. it is connected to the boost inductor and the boost diode. peak switch current is limited internally to 450ma. if an external v dd supply is used, leave this pin open. gnd (pin 5): device ground pin. this pin is ground for the boost switch, gate driver as well as the control circuitry. tie the v in and v dd bypass capacitors and ground plane close to this pin to minimize the effects of switching currents on part performance. typical sw pin waveform sw pin waveform at maximum boost regulator output v sw 5v/div v in = 5v 10 m s/div 3451 g12 4.7 m h inductor v sw 5v/div v in = 5v 10 m s/div 3451 g13 4.7 m h inductor typical perfor a ce characteristics uw specifications are at t a = 25 c unless otherwise specified.
lt4351 6 sn4351 4351fs uu u pi fu ctio s ov (pin 6): overvoltage shutdown pin. this pin is used for input overvoltage detection. it is connected to a resistive divider from v in . when the voltage exceeds the ov threshold (0.3v), gate is pulled to gnd disabling power transfer. in addition, the fault pin pulls low indicating a fault. overvoltage detection has filtering on it to prevent false triggering. the filtering depends on the level of overdrive. filtered tripping will occur when ov exceeds 0.3v. if ov exceeds 0.33v, the gate immediately turns off (no filtering). if overvoltage detection is not required, ground the ov pin. see applications informa- tion for further information. uv (pin 7): undervoltage shutdown pin. this pin is used for the undervoltage detect function. it is connected to a resistive divider from v in . when the voltage is below the uv threshold, gate pulls to gnd disabling power transfer. in addition, the fault pin pulls low indicating a fault. when the uv pin voltage drops below the threshold, a 10 m a current is pulled from the divider to provide hysteresis. if under- voltage detection is not required, tie the uv pin to a voltage greater than 320mv and not greater than v in . do not force more than 9v on uv due to an internal clamp. see appli- cations information for further information. fault (pin 8): fault comparator status pin. this pin pulls low when a fault occurs. a fault has occured if the uv pin is below threshold or the ov pin is above threshold. the fault pin low indicates that there is a problem with the v in (source) supply. gate is pulled to gnd during a fault, disabling the mosfet(s) and prohibits common supply contamination. if the gate pin goes to compliance (gate equals the lesser of v dd C 2.3v or out + 7.4v) and v in is greater than out by more than 0.21v, fault turns on as an indicator that the mosfets are probably not function- ing. leave this pin open if not used. status (pin 9): mosfet status pin. this pin pulls low when gate is above v in by more than 0.7v and v in is greater than out by 15mv. this indicates the mosfet is on. leave this pin open if not used. out (pin 10): common supply pin. this pin is connected to the supply common and is used in conjunction with v in as one input controlling the mosfet(s).
lt4351 7 sn4351 4351fs block diagra w + 15mv enable enable qsw c uv c ov 0.3v 0.3v 0.33v c ovf gnd ov uv r2 r1 r a sw v dd v in gate v in + + 3 2 4 6 5 7 open mosfet detect 600ns one shot + 1 10 9 8 10.7v reg + + r b + + v in from individual supply to common supply v out out out st status fault 4351 bd driver
lt4351 8 sn4351 4351fs operatio u increasingly, system designers have to deal with multiple supply sources. the multiplicity may provide parallel, re- dundant supplies for increased reliability or provide a means of connecting disparate supplies. in all cases the desire is for behavior like a diode but with no loss or voltage drop. oring diodes have been the conventional means of connecting these supplies. the disadvantage of this ap- proach is that diodes introduce efficiency loss because of their forward voltage drop. this variable voltage drop also degenerates supply tolerance. additionally, diodes pro- vide no information concerning the status of the sourcing supply. separate control must also be added to ensure that a supply that is out of range is not allowed to affect the common supply. the lt4351 eliminates these problems by using n-channel mosfets as the pass elements. the mosfet is turned on when power is being passed, allowing for a low voltage drop from the supply to the load. when the input source voltage drops below the output common supply voltage it turns off the mosfet, thereby matching the function and performance of an ideal diode. the lt4351 drives either a single mosfet or dual back- to-back mosfets. dual mosfets are chosen to eliminate current flow from the input supply to the output supply when the v in voltage is greater than out. a driver amplifier monitors the input (v in ) and output (out) and controls the mosfets. if v in exceeds out by 15mv, gate goes high and turns on the mosfet(s) allowing for power passage. undervoltage and overvoltage comparators c uv , c ov and c ovf also control power passage. a resistive divider in conjunction with the uv and ov pins sets appropriate thresholds such that the mosfet(s) is off when the uv pin is below 300mv or ov pin is above 300mv. to help deal with the transients on the supply lines, the uv input has current hysteresis. when the uv voltage drops below the 300mv threshold, a 10 m a current is pulled from the pin. thus the user can set the hysteresis level through appropriate values in the divider. overvoltage shutdown occurs in two stages. the first occurs when the ov pin exceeds the 300mv reference. when ov just exceeds the reference, an internal capacitor starts charging, delaying the signal to turn off the mosfet(s). the second occurs when the ov pin exceeds 330mv. the ovf comparator will immediately trip pulling gate to gnd. this affords a delay inversely proportional to the amount of overdrive. this also provides for glitch immu- nity without compromising response time in the event of a serious overvoltage condition. the fault output indicates the status of the c ov , c ovf and c uv comparators. it pulls low during a fault condition. it also pulls low when gate is at compliance and v in > out by more than 0.21v indicating a probable nonfunctioning mosfet. compliance occurs when gate is at the lesser of out + 7.4v or v dd C 2.3v. fault derives its drive from the greater of v in or out. it is active if v in or out is greater than 0.9v. if v in or out is below this level, the output state is not guaranteed. the gate drive consists of a high current, wide bandwidth amplifier (driver). when the amplifier is enabled, it at- tempts to regulate the gate voltage such that the voltage across the mosfet(s) is approximately 15mv. if the mosfet(s) on resistance is so high as to prevent regula- tion, then gate goes to compliance and the mosfet(s) fully turns on. the inputs to the amplifier are v in and out. the gate pin sources current from v dd and sinks current to gnd. the maximum gate to v in voltage is the lesser of v dd C 2.3v or 7.4v above v out or v in (internal clamp voltage). the status comparator, st, pulls low when gate ex- ceeds v in by 0.7v. this occurs when v in > out + 15mv. the status pin pulls low as an indication that power is passing through the mosfet(s). if v in is greater than out by 0.21v and gate > v in + 7.4v or at compliance (gate = v dd C 2.3v), status will go high as an indication of a likely open mosfet. fault will pull low in this state indicating the probable fault. the gate drive amplifier and status function derive power from v dd . the circuit requires v dd > 2.5v. if v dd is present, the gate drive amplifier and status are active independent of the state of v in . if in a fault, gate pulls
lt4351 9 sn4351 4351fs applicatio s i for atio wu uu setting fault thresholds the gate drive amplifier implements the ideal diode func- tion. the fault comparators (uv and ov) prevent out of range input voltages from affecting the output by disabling the amplifier during these conditions. think of the uv and ov as gating the ideal diode function, something a regular diode cant do. a resistive divider from v in to uv and one from v in to ov are the usual way of setting the fault thresholds. for uv the resistor values are set by: r uv i r v uv v r hyst uvhyst uv fault uv 2 12 = = where uv hyst is the desired undervoltage hysteresis at the input. uv fault is the desired undervoltage trip voltage at the input. v uv is the part undervoltage trip point (0.3v) and i hystuv is the undervoltage hysteresis current (10 m a). see figure 1. the divider on the ov pin is a straightforward resistive divider (figure 2): r ov v r r v r r divider current b fault ov a a ab = ? ? ? ? = . , 1 03 where ov fault is the desired overvoltage trip point at the input and v ov is the ov pin threshold (0.3v). the ov pin has 7mv of voltage hysteresis at room. it is possible to do both dividers together using only three resistors though with more interdependence in compo- nents (figure 3). the input bias current for uv and ov is less than 200na, so keep resistor values less than 10k. r2 r1 uv i hys 10 a v uv 300mv v in uv turning on uv turning off r2 r1 uv i hys 10 a v uv 300mv 4351 f01 v in figure 1 r b r a ov v ov 300mv 4351 f02 v in figure 2 r2 r3 r1 ov uv 4351 f03 v in figure 3 c1 r2a r2b r1 uv 4351 f04 v in figure 4 actively low. in the event of v dd collapse there still is an active pull-down (though of lesser strength) of gate powered from out, guaranteeing turn off. the on-chip boost regulator uses a constant off-time control scheme. when v dd is below the regulation trip voltage, the switch turns on after a 600ns off-time. when the switch turns on current ramps up in the inductor until the current limit is reached (450ma). the switch turns off and the inductors current flows through the external diode to charge up the v dd capacitor. if v dd is still too low, the switch turns on again after a fixed off-time of 600ns. the boost regulator regulates v dd to approximately 10.7v above v in . when v dd is above this level, the sw transistor turn-on is disabled. when v dd falls below this level by the hysteresis level, the sw transistor is allowed to turn on. there is approximately 0.15v of hysteresis. operatio u
lt4351 10 sn4351 4351fs applicatio s i for atio wu uu in that case the resistor values are set by: r uv i r v uv ov v uv v r r vuv ov uv v r hyst uvhyst uv fault fault ov fault uv ov fault fault fault uv 3 23 13 = = = () hysteresis helps prevent erratic behavior due to the noise on v in . two of the most common noise sources are: v in dipping when the mosfets first turn on and draw down the voltage on the v in capacitors, and the boost regulator switch turning on and drawing current from the v in capacitors. use low esr capacitors for v in and out filtering. note that because the uv pin uses current hysteresis, placing a capacitor on uv to ground to filter noise will reduce the effective hysteresis. filtering can be achieved by splitting the r2 resistor as shown in figure 4. to defeat undervoltage fault detection, the uv pin should be tied higher than 0.33v. uv can be tied to v in provided v in < 9v. overvoltage fault detection can be defeated by grounding the ov pin. do not exceed v in . boost regulator the boost regulator will start working as soon as v in is greater than 0.85v. the regulator will supply all the current for the gate drive amplifier. while the amplifier itself requires only about 3ma, larger current pulses are re- quired when charging the mosfet gate. the reservoir capacitor on v dd will provide this current (figure 6). overvoltage filtered fault input referred ov referred uv referred v uv = 0.33v v uv = 0.3v v uv < 0.3v v ov > 0.3v v ov = 0.3v 4351 f05 ov fault uv fault + uv hyst uv fault undervoltage hysteresis overvoltage fault: gate low undervoltage fault: gate low gate controlled by v in ?v out figure 5. graphical representation of the uv and ov functions figure 6 l1 d1 d2 qsw gnd sw v dd c dd 4351 f06 v in lt4351 external shutdown to externally turn off the mosfets, such as to disable the supply, use an open-collector transistor pulling down on the uv pin. note this will not turn off the boost regulator which will continue to operate. the regulator performance is relatively insensitive to the inductor value. the inductor value does control the fre- quency of operation. a 4.7 m h inductor is recommended for v in voltages less than 10v and 10 m h for v in voltages greater than 10v. several inductors that work well with the lt4351 are listed in table 1. many different sizes and shapes are available. consult each manufacturer for more detailed information and for their entire selection of related parts. the switching frequency for the boost regulator is around 1mhz so ferrite core inductors should be used to obtain the best efficiency. the inductor must handle a peak current of 0.7a minimum and have a dc resistance of 0.5 w or less. shielded inductors are recommended to reduce the noise due to inductive switching. table 1. recommended inductors max ind dcr part number ( m h) (m w ) vendor ds1608c-472 4.7 90 coilcraft ds1608c-103 10 160 (847) 639-6400 www.coilcraft.com cdrh4d18-4r7 4.7 125 sumida cdrh4d28-100 10 95 (847) 956-0666 www.sumida.com lmnp045b4r7n 4.7 50 taiyo yuden lmnp045b100m 10 52 (408) 573-4150 www.t-yuden.com
lt4351 11 sn4351 4351fs for v in less than 2v, choose a dc resistance less than 0.2 w . note that v dd current referred to the input supply is higher. a first order approximation of the input current is: i v i vinvdd in vdd =+ ? ? ? ? 1 10 6 80 . % under normal operation, the v dd current is under 10ma and the boost regulator operates in burst mode operation. if any additional load is added, you must ensure that the regulator is capable of supplying that load. as the load is increased, the boost regulator will switch into continuous mode operation. further increases in load will collapse the boost regulator voltage. operating the regulator with increased load will cause increased ic power dissipation and temperature, which must be taken into consideration. a 100ns delay from detecting the switch current limit to turning off the power switch produces an overshoot of the inductor current from the 0.45a switch limit. the amount of overshoot depends on the boost regulator inductance. choosing an inductor that can handle 0.75a peak current will be sufficient for the recommended inductors. diode selection schottky diodes, with their low forward voltage drop and fast switching speed, are the best match for the lt4351 boost regulator. select a diode that can handle 0.75a peak current and a reverse breakdown of 15v greater than the maximum v in . v dd capacitor selection low esr (equivalent series resistance) capacitors should be used on v dd to minimize the output ripple voltage. multilayer ceramic capacitors are the best choice, as they have a very low esr and are available in very small packages. always use a capacitor with a voltage rating at least 12v greater than v in . capacitors two types of input capacitors are generally needed for the lt4351. the first is a large bulk capacitor that takes care of ringing associated with inductance of the input supply lines and provides charge for the load when switching the mosfet. the input parasitic inductance in conjunction with c b and its esr create an lcr network. the input lcr can be stimulated by the boost regulator switch current or load current transients when the mosfets are on. to reduce ringing associated with input inductance, c b should be: c l r b in esr 3 4 2 where c b is the capacitor value, r esr is the capacitor?s esr and l in is the inductance of the input lines. while damped ringing is not necessarily bad, it may produce unexpected results as the lt4351 ideal diode reacts to the varying v in to out voltage. typically an electrolytic or tantalum low esr capacitor would be used. figure 7a illustrates v in for a low value of c b and figure 7b shows it with a correctly sized value. applicatio s i for atio wu uu figure 7a. example of input voltage ringing with low c in capacitor at mosfet turn off v in 200mv 10 m s/div 4351 f07a figure 7b. example of input voltage with sufficient c in capacitor at mosfet turn off v in 200mv 10 m s/div 4351 f07b
lt4351 12 sn4351 4351fs as an example, for 500nh of inductance and r esr of about 100m w , then: c nf f 3=m 4 500 01 200 2 . check vendor data for esr and iterate to get the best value. additional c b capacitance may be required for load concerns. if the boost regulator is being used, place a 10 m f low esr ceramic capacitor from v in to gnd. place a 10 m f and a 0.1 m f ceramic capacitor close to v in and gnd. these capacitors should have low esr (less than 10m w for the 10 m f and 40m w for the 0.1 m f). these capacitors help to eliminate problems associated with noise produced by the boost regulator. they are decoupled from the v in supply by a small 1 w resistor as shown in figure 8. the lt4351 will perform better with a small ceramic capacitor (10 m f) on out to gnd. back-to-back mosfets prevent the mosfet body diode from passing current. use a single mosfet if current flow is allowable from input to output when the input supply is above the output (limited overvoltage protection). in this case the mosfet should have a source on the input side so the body diode conducts current to the load. back-to-back mosfets are normally connected with their sources tied together to provide added protection against exceeding maximum gate to source voltage. selection of mosfets should be based on r ds(on) , bv dss and bv gss . bv dss should be high enough to prevent breakdown when v in or out are at their maximum value. r ds(on) should be selected to keep within the mosfet power rating at the maximum load current (i 2 ? r ds(on) ) bv gss should be at least 8v. the lt4351 will clamp the gate to 7.5v above the lesser of v in or out. for back-to- back mosfets where sources are tied together, this allows the use of mosfets with a vgs max rating of 8v or more. if a single mosfet is used, care must be taken to ensure the vgs max rating is not exceeded. when the mosfet is turned off, the gate voltage is near ground, the source at v in . thus, mosfet vgs max must be greater than v in(max) . if a single mosfet is used with source to v in , then bv gss should be greater than the maximum v in since the mosfet gate is at 0.2v when off. the gate drive amplifier will attempt to regulate the voltage across the mosfets to 15mv. regulation will be achieved if: r mv i ds load < 15 2 for two mosfets and r mv i ds load < 15 for a single mosfet this requires very low r ds values. this may be achieved by paralleling mosfets, but be careful to keep intercon- nection trace resistance low. in the event that regulation cannot be achieved, the gate drive amplifier will drive gate to its clamp and achieve the best r ds possible at that level. applicatio s i for atio wu uu figure 8. v in capacitors external boost supply the v dd pin may be powered by an external supply. in this case, simply omit the boost regulator inductor and diode and leave the sw pin open. suitable v dd capacitance (minimum of a 1 m f ceramic) should remain due to the cur- rent pulses required for the gate driver. the v dd current consists of 3.5ma of dc current with the current required to charge the mosfets gate which is de- pendent on the gate charge required and frequency of switching. typically the average current will be under 10ma. mosfet selection the lt4351 uses either a single n-channel mosfet or back-to-back n-channel mosfets as the pass element. v in 1 c v3 10 f c v1 10 f c b l in parasitic c v2 0.1 f v in gate lt4351 4351 f08 gnd
lt4351 13 sn4351 4351fs status the status pin sinks current when the input (v in ) is above output (out) by 15mv and gate is above v in by 0.7v. this will normally indicate that power is being passed though the mosfets. in the event of a nonfunctional mosfet, the gate voltage will be driven high (to the gate clamp voltage). if v in is greater than out by more than 0.21v, the fault pin will sink current to signal the potential problem. there is no direct measurement or confirmation of current flowing in the mosfets. current is shared between sources based on their voltage and series resistance. if precision load sharing is desired, the ltc4350 may be a more suitable part. redundant supplies the lt4351 is an improved solution for oring redundant supplies because of its lower forward drop versus conven- tional diodes. the lower forward drop significantly im- proves overall efficiency, improves the voltage tolerance at the load and provides for a more accurate transition from supply to supply and more accurate load sharing between supplies. applicatio s i for atio wu uu oring can be done either at the load or at the source. figure 9 shows some examples. oring at the load is usually the safest method since it protects against shorts in interconnects. the lt4351 tighter forward voltage tolerance makes it easier to balance current between similar supplies using the droop method. the droop method uses the supply voltage and series resistance in the power path to provide load sharing. in this case, size the mosfets r ds(on) low to allow for regulation. oring disparate supplies the lt4351 provides an easy solution for connecting together different types of power sources. again because of the low forward drop, the efficiency of the system is improved and the voltage transition between supplies is more accurate. in addition, the undervoltage and overvolt- age features of the lt4351 provide options for enabling and disabling the supplies that are not available from a common diode. figure 10 shows some examples of con- necting disparate supplies. figure 9. redundant backplane supplies figure 10 lt4351 board lt4351 load source 1 backplane source 2 lt4351 board lt4351 load 4351 f09 source 1 backplane source 2 lt4351 isolated system supply from wall adapter isolated battery backup three source oring provides protection against out of range supplies wall adapter system supply load lt4351 battery wall adapter load + lt4351 battery lt4351 lt4351 wall adapter load 4351 f10 + system supply
lt4351 14 sn4351 4351fs applicatio s i for atio wu uu start-up considerations there is no inherent shutdown in the part. as v in ramps up, the boost regulator starts at about 0.85v and becomes fully operational by 1.1v. the undervoltage and overvoltage comparators become accurate by 1.2v. the gate drive amplifier keeps gate low during this period with either a passive pull-down, a weak active pull-down if out is greater than 0.8v or with the full gate drive sink if v dd is above 2.2v. once v in is greater than 1.2v and v dd is up, the part then operates normally. the uv and ov pins will control the enabling of the gate driver and once enabled, the v in to out voltage controls mosfet turn on. if v dd is still being charged when the gate driver turns on the mosfet, the gate pin tracks with the v dd increase until it reaches either the gate clamp voltage or the compliance of the gate driver. if v dd is present without v in or out, the gate pin actively sinks low. power dissipation the internal power dissipation of the lt4351 is comprised of the following four major components: dc power dissi- pation from v in , dc power dissipation from v dd , the dissipation in the boost switch including the base drive, and dynamic power dissipation due to current used to charge and discharge the mosfets. the dc components are: p dcvin = i vin ? v in p dcvdd = i vdd ? v dd figure 11 shows the internal dissipation of the boost regulator as a function of v in and inductor value. figure 11 represents the worst-case condition with the regulator on all the time, which does not occur in normal practice. since the boost regulator supplies current for v dd , the current is the v dd supply current (3.5ma) plus the average current to charge the gate. for a gate charge of 50nc at a 10khz rate, this adds 0.5ma of current. the power dissi- pated by the boost regulator to supply the 4ma is shown in figure 12, representing a more typical situation. finally, the gate driver dissipates power internally when charging and discharging the gate of the mosfets. this power depends on the input capacitance of the mosfets and the frequency of charge and discharge. the power associated with this can be approximated by: pfvq v gate g dd g in = ? ? ? ? 1 16 where q g is the required gate charge to charge the mosfet to the clamp voltage (7.4v) and f g is the fre- quency at which the gate is charged and discharged. normally f g is low and the resulting power would be very low. figure 13 shows p gate for a 50nc gate charge at a 1khz rate. total power dissipation is the sum of all of p dcvin , p dcvdd , p boost and p gate . figure 14 is representative of the total power dissipation of a typical application at steady state. figure 11. p boost(max) figure 12. p boost(typ) v in (v) 0 p boost (w) 0.20 0.25 l = 10 h l = 4.7 h 20 4351 f11 0.15 0.10 5 10 15 0.30 v in (v) 0 p boost (w) 0.015 0.020 l = 4.7 h 20 4351 f12 0.010 0.005 5 10 15 0.025
lt4351 15 sn4351 4351fs applicatio s i for atio wu uu the die junction temperature is then computed as: t j = t a + q ja ? p total where t j is the die junction temperature, t a is the ambient temperature, q ja is the thermal resistance of the part (120 c/w) and p total is ascertained from the above. therefore, a 0.1w power dissipation causes a 12 degree temperature rise above ambient. design example the following demonstrates the calculations involved for setting design components for a 5v system that requires 5a. two supplies are used to do this. the v in supply will be deemed in spec when it is within 5% of nominal. allow 5% of hysteresis for uv. so, uv fault = 4.75v, uv hyst = 0.25v ov fault = 5.5 two separate resistive dividers are used. for the uv divider: r uv i v a k use k r rv uv v kv vv hyst uvhyst uv fault uv 2 025 10 25 24 9 1 2 249 03 475 03 == m = () == . . .. .. r1 = 1.68k. the closest 1% value is 1.69k figure 14. total power (typical) figure 13. p gate vs v in (v dd = v in + 10.7) the ov resistors are set as a straight resistive divider. if the current in the ra, rb divider is 200 m a, then: r r ov v k a b fault ov = m = = ? ? ? ? = ? ? ? ? = 0.3v 200 a 1.5k use 1.47k (1%) then r r 25.48k use 25.5k a b . . . 1 55 03 1147 for regulation, the mosfets must have: r mv a m ds <=w 15 25 15 . this very low value cannot be accomplished with a single set of mosfets so a decision must be made whether to use multiple mosfets or to live with an unregulated offset. since low m w r ds(on) is available, the ir drop using a single fet would still be acceptable. for r ds(on) = 4m w the drop is 2 ? 5a ? 4m w = 40mv. the finished schematic is shown in figure 15. v in (v) 0 p gate (w) 0.002 0.003 20 4351 f13 0.001 0 5 10 15 0.004 f gate = 1khz q g = 50nc v in (v) 0 power (w) 0.10 0.12 20 4351 f14 0.08 0.06 5 10 15 0.16 0.14 l = 10 h l = 4.7 h v dd = v in + 10 0.5ma gate current
lt4351 16 sn4351 4351fs figure 15. 5v/5a design example layout considerations there are two considerations for board layout. the first is that v in and v dd bypass capacitors should be as close to the part as possible. the gnd pin should represent the common tie point. the resistive dividers for uv and ov should tie here as well. applicatio s i for atio wu uu take care that current flow to the load (both through v in and gnd), does not inadvertently produce errors due to i r drops in pcb traces. keep the traces to the mosfets wide and short and close to the part. the pcb traces associated with the power path through the mosfets should have low resistance. gate out v in uv ov 1 f 10 f mbr0530 mbr0530 r1 1.69k 1% r b 25.5k 1% v in 5v 4.7 h lt4351 sw v dd status 5 fault gnd si4838dy 3 7 6 4 2 110 9 8 r a 1.47k 1% r2 24.9k 1% 1 0.1 f 220 f 10 f 5v 2k 4351 f15 out 2k 10 f
lt4351 17 sn4351 4351fs typical applicatio s u gate out v in uv ov 1 f 10 f mbr0530 mbr0530 r1 365 1% r b 73.2k 1% 10 h 12v lead-acid battery charger lt4351 sw v dd status 5 fault gnd si4408dy 3 7 6 4 2 110 9 8 r a 1.5k 1% r2 12.7k 1% 1 0.1 f 220 f 10 f 10 f 10k 4351ta02 out load 10k uv fault = 10.8v ov fault = 15v 14v power supply + + lead acid battery backup
lt4351 18 sn4351 4351fs typical applicatio s u gate out v in uv ov 1 f r1 1.69k 1% r b 25.5k 1% lt4351 sw v dd status 5 fault gnd 1st source si4838dy 3 1 10 f 7 6 4 2 110 9 8 r a 1.47k 1% r2 24.9k 1% 0.1 f 100 f 10 f 10 f 2k 4351 f15 2k load 4351 ta03 common 5v source 12v source 2nd 5v source 2nd 12v source 2nd lt4351 circuit + 5v redundant supply with external v dd
lt4351 19 sn4351 4351fs u package descriptio ms package 10-lead plastic msop (reference ltc dwg # 05-08-1661) msop (ms) 0603 0.53 0.152 (.021 .006) seating plane 0.18 (.007) 1.10 (.043) max 0.17 0.27 (.007 ?.011) typ 0.127 0.076 (.005 .003) 0.86 (.034) ref 0.50 (.0197) bsc 12 3 45 4.90 0.152 (.193 .006) 0.497 0.076 (.0196 .003) ref 8 9 10 7 6 3.00 0.102 (.118 .004) (note 3) 3.00 0.102 (.118 .004) (note 4) note: 1. dimensions in millimeter/(inch) 2. drawing not to scale 3. dimension does not include mold flash, protrusions or gate burrs. mold flash, protrusions or gate burrs shall not exceed 0.152mm (.006") per side 4. dimension does not include interlead flash or protrusions. interlead flash or protrusions shall not exceed 0.152mm (.006") per side 5. lead coplanarity (bottom of leads after forming) shall be 0.102mm (.004") max 0.254 (.010) 0 ?6 typ detail ? detail ? gauge plane 5.23 (.206) min 3.20 ?3.45 (.126 ?.136) 0.889 0.127 (.035 .005) recommended solder pad layout 0.305 0.038 (.0120 .0015) typ 0.50 (.0197) bsc information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
lt4351 20 sn4351 4351fs linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 l fax: (408) 434-0507 l www.linear.com ? linear technology corporation 2003 lt/tp 1203 1k ? printed in usa part number description comments ltc1473 dual powerpath tm switch driver switches and isolates sources up to 30v ltc4350 hot swappable load sharing controller output voltage 1.2v to 12v, equal load sharing ltc4412 low loss powerpath controller in thinsot tm p-channel mosfet, 3v to 28v range powerpath and thinsot are trademarks of linear technology corporation. related parts typical applicatio u gate out v in uv ov 1 f mbr0530 mbr0530 r1 1.07k 1% 10 h 12.6v battery1 charger lt4351 sw v dd status 5 fault gnd si4408dy 3 1 1 7 6 4 2 110 9 8 r2 40.1k 1% 0.1 f 10 f 100 f uv fault = 11.8v 10 f 10k 5% 10k 5% out 100k 5% 10 f + gate out v in uv ov 1 f mbr0530 mbr0530 300k 5% 10 h 1n914 12.6v battery2 charger lt4351 sw v dd status 5 fault gnd power is switched to battery2 when battery1 drops to 11.8v si4408dy 3 7 6 4 2 110 9 8 120k 5% 10 f 0.1 f 10 f 10 f 10k 4351ta04 out 100 f 10k load + + 100 f + primary battery with secondary battery backup

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